1. Field of the Invention
The invention relates to an oscillator, especially for microwave circuits.
2. Discussion of the Background
An oscillator is known, for example, from US 2005/0270114 A1.
The phase noise of a voltage-controlled oscillator has a significant influence on the quality of the output signal of the system, in which the oscillator is used. It is therefore important in designing voltage-controlled oscillators to suppress the power of the phase noise relative to the useful signal as far as possible. Amongst other influences, the quality factor of the resonator, the bandwidth of the resonator and an optimum start-up reserve of the oscillator play a significant role with regard to the phase noise.
Tuneable oscillators for the microwave range conventionally provide a resonator, which can be tuned with a varactor diode and an amplifier coupled to the latter, also referred to below as the oscillator core. Bipolar transistors are used as the amplification element in a common-base or common-emitter configuration. Instead of bipolar transistors, field-effect transistors can also be used in a common-gate or common-source configuration.
By way of explanation of the problem, upon which the present invention is based, FIG. 1 shows a hitherto conventional oscillator 1 consisting of a resonator 2 and an amplifier 3, which is also referred to as the “core”. The resonator 2 and the amplifier 3 are connected to one another at the coupling position 4. The central component of the resonator 2 is a tuneable element 5, which is conventionally formed by a varactor diode, also referred to as a variable-capacitance diode. Applying an electrical voltage to the varactor diode in the reverse direction changes the length of the space-charge zone, and therefore also the capacitance of the varactor diode. For this purpose, the anode of the varactor diode is connected, in the exemplary embodiment via an inductance 6, to the circuit ground 7, and the cathode of the varactor diode 5 is connected via a choke 8 to a tuning voltage, which is positive relative to the ground 7, and which is described in FIG. 1 as the “varactor bias”.
As an alternative, it is, of course, also possible, to attach the cathode of the varactor diode 5 to the circuit ground and to apply a voltage, which is negative relative to the ground, to the anode. The cathode of the varactor diode 5 is connected via a series capacitor 9 to the circuit ground 7. The capacitance of the varactor diode 5 and the capacitance C2 of the capacitor 9, together with that of the inductance 6 with the inductance value L3, form a resonant oscillating circuit.
The amplifier 3 comprises an amplification element 9. In the exemplary embodiment presented, the amplification element 9 is a bipolar transistor, which is operated in common-base configuration, that is to say, with the base connected to the circuit ground. As an alternative, a common-emitter configuration can also be considered within the framework of the present application. A field-effect transistor, preferably in common-gate configuration, can be used instead of a bipolar transistor. In this context, a common-source configuration can also be considered instead of the common gate-configuration.
In the exemplary embodiment illustrated in FIG. 1, the bipolar transistor 9 is connected by the emitter end to the coupling position 4. The collector of the transistor 9 is connected via a choke 10 to the collector voltage “collector bias”. At the same time, the high-frequency output voltage RF Out can be tapped at the collector of the transistor 9 via a capacitor 11. The base voltage “base bias” is supplied to the base via a choke 12 and an inductance 13. The node between the inductance 13 and the choke 12 is connected via a capacitor 14 to the circuit ground 7.
In order to operate the oscillator 1 according to FIG. 1, two conditions must be fulfilled in the steady state of the oscillator 1 as described below.
Initially, in the steady state, the loop amplification of the resonator 2 and the amplifier (core) 3 must be equal to 1, that is to say, the amplifier 3 must compensate the losses of the resonator 2. In other words, the quotient of the real part of the complex resistance of the amplifier viewed from the coupling position 4 and of the real part of the complex resistance of the resonator 2, also viewed from the coupling position 4 must be −1. The complex resistance of the amplifier (core) Z_Core, which is obtained on looking from the coupling position 4 towards the amplifier 3, and the complex resistance Z_Resonator, which is obtained on looking from the coupling position 4 towards the resonator 2, are visualized respectively in FIG. 1. The fact that the real part of the complex resistance Z-Core of the amplifier (core) is negative, results from the amplification (negative resistance). However, for a secure start-up of the oscillator 1, under small signal condition, it must also be ensured that an adequate start-up reserve is available, that is to say, that the quotient is significantly smaller (of a larger value) than −1, ideally approximately −3.
The second condition is a phase condition and leads to the situation that the sum of the imaginary part of the complex resistance of the resonator Z_Resonator and of the amplifier Z_Core at the resonant frequency must be equal to 0. Considered in visual terms, this means that a wave travelling from the coupling position 4 to the resonator 2, where it is reflected, and then travelling to the amplifier 3, where it also reflected with amplification, and finally returning to the coupling position 4, may have changed its phase only by an integer multiple of 2π (positive feedback), so that the system is resonant for this frequency.
FIG. 2A presents the imaginary part of the sum of the two complex resistances Z_Resonator and Z_Core for a given tuning voltage of the varactor diode 5. It is evident that the second condition Im(Z_Resonator)+Im(Z_Core)=0 at 4.75 GHz is fulfilled. As shown in FIG. 2B, at the same frequency, a ratio of −3.381 is obtained for the quotients of the real parts of the two complex resistances, thereby providing an adequate start-up reserve.
However, the oscillator 1 must be operated over a relatively wide frequency range, wherein the resonant frequency should be variable over several GHz. FIG. 3A presents the real part, and FIG. 3B presents the imaginary part of the complex resistance Z_Core of the amplifier 3. For the example presented, a resonance, which significantly determines and limits the tuning bandwidth is recognizable at approximately 6 GHz. This is evident when adjusting the tuning voltage of the varactor diode 5 of the resonator 2.
FIGS. 4A, 4C and 4E present the imaginary parts of the sum of the imaginary parts of the complex resistances Z_Resonator and Z_Core for different tuning voltages of the varactor diode of 0 V, 12.5 and 25 V. FIGS. 4B, 4D and 4F correspondingly present the quotient of the real parts for the same tuning voltages. It is evident that the second condition for a diminishing imaginary part can be achieved everywhere; however, that the first condition can no longer be achieved everywhere. For example, it is no longer achieved in FIG. 4F for the tuning voltage 25 V; the quotient is even positive. In FIG. 4D, a quotient of only −2.010 is achieved for the tuning voltage 12.5 V; in this context, an adequate start-up reserve should still be available. The invention is therefore a response to the problem that the conditions for operating the oscillator according to the prior art can be achieved only over a relatively-limited frequency range.